OVER THE last decade, there has been growing interest

Transcription

1 314 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 47, NO. 1, JANUARY/FEBRUARY 2011 A New Low-Cost Hybrid Switched Reluctance Motor for Adjustable-Speed Pump Applications Kaiyuan Lu, Peter Omand Rasmussen, Steve J. Watkins, Member, IEEE, and Frede Blaabjerg, Fellow, IEEE Abstract This paper presents a new low-cost hybrid switched reluctance (HSR) motor intended for use in adjustable-speed pump drive systems. The motor is a single-phase motor, driven by a unipolar converter, which uses both the reluctance torque and the permanent magnet interaction torque. Compared with conventional single-phase switched reluctance motors, it has an increased torque density. The cogging torque is beneficially used in this motor for reducing the torque ripple. It is demonstrated that such a motor drive system can be a suitable candidate to advantageously compete with the existing motor drive systems for low-cost applications. Finite-element models are used to analyze and predict the motor s performance. The proposed motor drive system has been fabricated, and its performance has been tested in the laboratory. These experimental results are also presented. Index Terms Adjustable-speed, hybrid switched reluctance motor, low-cost. I. INTRODUCTION OVER THE last decade, there has been growing interest in new adjustable-speed switched reluctance (SR) and permanent magnet (PM) drives to replace the conventional induction motor drives for applications like pumps. In these applications, the cost constraint has dictated the need for lowcost motors and drive systems. For the pumps to be used indoors, acoustic noise and efficiency are considered to be important factors. To reduce the cost of a drive system, one of the options is to use a drive system with a reduced number of power electronic switches. Less power devices would also simplify the control strategy and drive circuitry, which increase the robustness of the drive system. Often, bipolar converters require more power switches than unipolar converters. The cost of a drive system could be reduced by substituting the bipolar converters with unipolar converters controlling the same type of motors. For example, in [1] and [2], two low-cost unipolar drive systems were introduced for controlling a three-phase brushless dc motor. Another possible Manuscript received January 15, 2007; accepted April 25, Date of publication November 9, 2010; date of current version January 19, Paper MSDAD , presented at the 2006 Industry Applications Society Annual Meeting, Tampa, FL, October 8 12, and approved for publication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the Appliance Industry Committee of the IEEE Industry Applications Society. K. Lu, P. O. Rasmussen, and F. Blaabjerg are with the Institute of Energy Technology, Aalborg University, 9220 Aalborg, Denmark ( klu@iet.aau.dk; por@iet.aau.dk; fbl@iet.aau.dk). S. J. Watkins is with Fleadh Electronics Ltd, Leeds, LS6 2RU, U.K. Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /TIA method to reduce the number of power switches and, thus, the cost of the drive system is to reduce the number of phases of the motor, without sacrificing the motor s performance. The simplest motor topology would be a single-phase motor, controlled by a unipolar converter, and these are of great interest for low-cost applications. In [3], Muller presented a single-phase motor with a PM rotor and variable air-gap length for low-cost applications. This motor is simple, cheap, and is able to self-start. However, it needs bipolar excitation, and its torque will fall to zero twice in one rotational period. The torque ripple is high. A similar design was reported in [4], where the author used a notched rotor to solve the self-starting problem. The single-phase PM motor may be also constructed in a transverse flux manner, as presented in [5], [6]. This kind of motors may only be suitable for the loads like fans, due to the high torque ripple problem. A PM motor naturally requires bipolar currents, and even for a single-phase PM motor, it may need four switches. If unipolar converters are used, the PM motor may only be able to produce half of the torque due to half-wave operation. To achieve the same output torque as controlled by bipolar converters within the same motor size, the unipolar current needs to be increased, which results in increased copper loss and reduced motor efficiency. The PM motor may also use a bifilar winding to accommodate the unipolar converter. However, a bifilar winding requires much more slot area. The copper loss of a bifilar winding will be high, and it may be difficult to improve efficiency. In short, PM motors controlled by unipolar converters would result in poor utilization of the motor. Unlike the PM motors, SR motors suit naturally unipolar excitations and are more interested for low-cost applications. There has been a lot of focus on minimizing the number of switching devices to reduce the drive cost for SR motors. For example, a single-controllable-switch converter [7] was proposed for controlling an asymmetric two-phase SR motor, and a modified topology was proposed in [8] for controlling the same type of motor with improved controllability of the current. A comparison of different converter topologies for lowcost multiphase SR motor drives was presented in [9]. Aside from the modification on the converter side to reduce the cost, the use of single-phase SR motors will be another good choice to achieve a low-cost solution. In [10], Compter introduced a low-cost drive system using a single-phase SR motor with two parking magnets. Similar motor designs could be found in [11], [12], where only one parking magnet was used. In addition to using the parking magnets, controlled saturation may be used to solve the self-starting problem of a single-phase SR motor, like the motors introduced in [13] and [14]. The rotors of these /$ IEEE

2 LU et al.: NEW LOW-COST HSR MOTOR FOR ADJUSTABLE-SPEED PUMP APPLICATIONS 315 TABLE I MAIN DIMENSIONS AND SPECIFICATIONS OF THE PROTOTYPE Fig. 1. Basic structure of the proposed single-phase HSR motor. motors have only two poles. It may be advantageous to design the motor with more rotor poles to increase its torque density, like the motor introduced in [15]. All these proposed singlephase SR motors suffer from a high torque ripple problem. There is no output torque during half of one electrical period of these single-phase SR motors. The Cyrano motor, presented in [16], uses the cogging torque as positive output torque, while the reluctance torque is zero, achieved by placing the parking magnets in the middle of two adjacent reluctance poles. To achieve a high positive cogging torque, thick magnets with high remanence flux densities are needed. These kinds of magnets are expensive and may not be suitable for low-cost applications. In this paper, a new single-phase hybrid switched reluctance (HSR) motor is presented. Like the Cyrano motor, this motor uses both the reluctance torque and the permanent magnet interaction torque. The cogging torque is beneficially used to reduce the torque ripple. Unlike the Cyrano motor, this motor uses cheap ferrite magnets assembled in a flux concentration manner to achieve a high positive cogging torque and offers more flexibility in shaping the cogging torque characteristic. The rotor of this motor is also particularly designed to reduce hydraulic loss. Finite-element (FE) models (FEMs) are used to analyze and predict the motor s performance, and the results are presented. This motor is controlled by a two-switch asymmetrical half-bridge converter. The experimental results for the prototype motor drive system are also given. The topology and working principle of the proposed HSR motor are presented in Section II. The measured back EMF, flux-linkage profiles, cogging torque, and interaction torque of the prototype motor are compared with the calculated results and presented in Section III. The power converter for the lowcost drive system is introduced in Section IV. The measured steady-state performance of the HSR motor drive system using a fan as the load is given in Section V. II. PROPOSED SINGLE-PHASE HSR MOTOR The proposed single-phase HSR motor is shown in Fig. 1. Some of the main dimensions and specifications of the motor are given in Table I. Fig. 2. Cogging torque and the total torque for a fixed dc current of the singlephase HSR motor obtained from 2-D FEM. This motor has four reluctance poles and two PM poles. There are four rectangular permanent magnets assembled in a flux concentration manner. The rotor pole has a small slot close to its surface, which functions like a stepped rotor pole providing the self-starting capability. The motor is designed to rotate in only the anticlockwise direction as shown in Fig. 1. The working principle of this motor could be interpreted as follows. When the rotor is in a position where the nonslotted parts of the rotor poles are aligned with the PM poles, if a demagnetization current, as shown in Fig. 1, is provided, the rotor will be pulled to the position aligned with the reluctance poles by a positive reluctance torque and a positive PM interaction torque. During this period, the cogging torque is negative. The current should be zero when the rotor has reached the position aligned with the reluctance poles. A positive cogging torque will then continue to pull the rotor poles to the position aligned with the PM poles. Repeating this procedure, a steadystate operation can be achieved. Fig. 2 shows an example of the instantaneous cogging torque and the instantaneous total torque for a fixed dc current for one electrical period. The results are obtained from a 2-D FEM. The position of 0 in Fig. 2 is the position shown in Fig. 1, where the center line of the whole rotor pole is aligned with the center line of the stator PM pole. It can be observed from Fig. 2 that the rotor will settle at a position near point A before starting. Positive torque will appear if the winding current is increased and the motor is able to selfstart. In steady-state operations, the current should be flowing between rotor angular positions B and C. After the rotor has

3 316 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 47, NO. 1, JANUARY/FEBRUARY 2011 Fig. 3. Demonstration of how the PM flux linkage could be used to increase the total torque. Fig. 5. Different cogging torque characteristics for different PM-pole arcs. Fig. 4. Illustration of the principle of flux concentration configuration. passed position C, the current should be zero, and the cogging torque will provide positive motoring torque. Like the Cyrano motor [16], by placing the PM pole in the middle of two adjacent reluctance poles, the torque density of this motor could be increased compared to conventional singlephase SR motors. This is shown in Fig. 3. For a normal SR motor, the average torque for a dc current is determined by the area enclosed by curve 1, curve 2, and line AB. If curve 2 could be moved along the negative fluxlinkage axis, like curve 3, the total enclosed area for the same current will be increased. The flux produced by the permanent magnets placed in the middle of the reluctance poles can be used to move the flux-linkage curve in such a manner as to increase the motor s torque density [16]. This increase is due to the additional PM interaction torque component. Unlike the Cyrano motor, by using the flux concentration structure, as shown in Fig. 4, low-cost magnets can produce a reasonably high air-gap flux density and, consequently, high positive cogging torque, which is the motoring torque when the current is zero. The flux concentration structure also offers more flexibility in shaping the cogging torque characteristic, e.g., by changing the width of the PM-pole arc, using nonuniform air gap under the PM pole, or using controlled saturation for the PM pole. Fig. 5 shows two different static torque characteristics for PM-pole arcs of 28 and 35.Itisshown in Fig. 5 that by reducing the PM-pole arc by 20%, the peak cogging torque value could be increased by about 35%. Fig. 6. Picture of the stator of the prototype HSR motor. It should be pointed out that the cogging torque has no contribution to the average torque. The cogging torque is negative in the region where the reluctance torque and the PM interaction torque are positive. The cogging torque is used to move some of the positive torque produced by the winding current to the region where the current is zero. Compared to a conventional single-phase SR motor, whose output torque is zero when the phase current is off, the torque ripple of this HSR motor is significantly reduced. The ideal cogging torque waveform should be trapezoidal and have an average positive value which equals the rated torque. The two key features, using low-cost ferrite magnets and having flexibility in shaping the cogging torque of the proposed HSR motor, advantageously compete with the existing Cyrano motor. A prototype of the proposed HSR motor has been manufactured. The stator and the rotor of the prototype are shown in Figs. 6 and 7. The rotor pole of the prototype is wider than the stator pole and has a 3-mm deep slot. When the motor is used as a pump motor, the rotor will be sitting in the water. By having the slot inside the rotor, which functions like a step, the rotor surface is smooth, and the hydraulic loss is reduced. The empty areas between the rotor poles are filled with special concrete, for reducing the hydraulic loss when the rotor is rotating in the water.

4 LU et al.: NEW LOW-COST HSR MOTOR FOR ADJUSTABLE-SPEED PUMP APPLICATIONS 317 Fig. 7. Picture of the rotor of the prototype HSR motor. Fig. 9. Comparison of the measured and calculated back-emf waveforms at a speed of 48 r/min. Fig. 8. Static test bench. III. STATIC MEASUREMENT RESULTS The test bench used to measure the prototype motor s back EMF, flux-linkage profiles, cogging torque, and interaction torque at different rotor positions is shown in Fig. 8. The stepping motor can rotate the test motor to a desired position and lock it. The power supply can provide ac, dc, and ac+dc energization voltages to the test motor for static torque and flux-linkage measurements. The measured torque, current, and voltage will be transferred to the computer via data acquisition cards. The measured and calculated back-emf waveforms are shown in Fig. 9. The results calculated are from 3-D FE analysis. The flux-linkage profiles are shown in Fig. 10, and the cogging torque waveforms are shown in Fig. 11. The measured interaction torque is shown in Fig. 12 for a fixed dc current of 1.38 A. The interaction torque is found by subtracting the measured cogging torque from the total torque with dc current energization. All the calculated results are obtained from a 3-D FEM. The shapes of the calculated and measured back-emf waveforms look similar, but the measured peak-to-peak value is about 30% lower than the calculated value. The measured minimum and maximum inductances are lower than the expected values, by 10% and 20%, respectively. These errors may be due Fig. 10. Comparison of the measured and calculated flux-linkage profiles at the minimum and maximum inductances positions. Fig. 11. Comparison of the measured and calculated cogging torque waveforms of the prototype HSR motor. to wrong air-gap lengths under the PM poles and the reluctance poles. Since the torque amplitude is small, the measurement accuracy is affected by the frictional torque, the misalignment

5 318 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 47, NO. 1, JANUARY/FEBRUARY 2011 Fig. 12. Comparison of the measured and calculated interaction torque, which is the total torque for a fixed dc current subtracted by the measured cogging torque. TABLE II MEASURED AND CALCULATED MAXIMUM AND MINIMUM INDUCTANCES AND THE CHANGE OF PM FLUX LINKAGES of the shafts of the torque transducer, the coupling, and the test motor, which may introduce a dc torque offset into the measured torque. This dc torque offset was found by averaging the cogging for 90 mechanical degrees and was removed. Using the measured instantaneous interaction torque, the average torque produced by the motor can be calculated. It should be noted that the cogging torque has no contribution to the average torque. The average torque may also be calculated using the measured minimum and maximum inductances and back EMF. Referring to Fig. 3, the average interaction torque equals T ave =4 Area BCDE 2π where 4 is the number of rotor poles, which is also the number of torque periods per 360 mechanical degrees. Assuming linear minimum and maximum inductance profiles, it can be approximated to T ave = 2 [ ] 1 π 2 I2 a(l max L min )+I a Δλ pm (2) where I a is the dc phase current. L max and L min are the maximum and minimum inductances. Δλ pm is the change of the PM flux linkage from minimum inductance position to maximum inductance position. The measured maximum and minimum inductances and Δλ pm are listed in Table II. The calculated values are also included for comparison. (1) Fig. 13. Average torque versus amplitude of square current waveform with constant amplitude between rotor angular positions at 0 and 45. The average torque versus current profile, calculated using (2) and the measured parameters, is shown in Fig. 13. The current has a square waveform with constant amplitude between rotor angular positions 0 and 45. The average torque calculated by averaging the positive torque part of measured instantaneous interaction torque waveform is also shown in Fig. 13 for comparison. These two curves agree with each other very well, which indicates reliable torque and inductance measurements. The measured average torque is about 30% lower than the expected value for the same winding current. This is due to the reduction in the minimum and maximum inductances and Δλ pm. To achieve the rated torque, the winding current has to be increased by 30%, and the copper loss would be 70% more than the designed value. IV. LOW-COST DRIVE ELECTRONICS In common with conventional doubly salient SR machines, the HSR motor requires only unipolar phase current for complete operation. In its simplest form, the power converter has only one controlled power switch to provide a path for phase energization and one diode to allow deenergization of a bifilar wound machine. However, the bifilar motor and power converter suffer from a number of drawbacks associated with leakage inductance stored energy loss and poorer winding utilization. There is also very little cost benefit in selecting a single-switch bifilar power converter due to the higher cost of 1000-V MOSFETS compared to 500-V devices. For this particular drive, a two-transistor power converter has been adopted, which is also inherently fault tolerant since there is no shootthrough path. Fig. 14 shows the two-transistor asymmetric bridge power-converter circuit. Full supply voltage is applied to the phase winding by firing the two MOSFETs simultaneously. Energy is returned to the dc link from the machine when both transistors are turned off and both diodes are conducting, clamping the voltage across the phase winding to negative dc-link voltage. Another important mode of operation occurs when only one transistor is turned on, which allows the phase current to freewheel in either the

6 LU et al.: NEW LOW-COST HSR MOTOR FOR ADJUSTABLE-SPEED PUMP APPLICATIONS 319 Fig. 14. Two-transistor asymmetric bridge power-converter circuit. high-side connected devices (Q1 and D1) or the low-side connected devices (Q2 and D2). This zero-voltage phase energization mode is advantageous in reducing switching losses in power electronics and can play an important role in current profiling for acoustic noise reduction. The peak current rating of the machine means that low-cost TO-220 packaged MOSFETs have been used (IRF840, 500 V), which offer low-loss high-efficiency performance. The drive only operates in the motoring quadrant so small axial-leaded diodes have been used (BYV26-600). The gate drive circuitry is a single independent low- and high-side gate driver IC with a boot-strap high-side supply generation (IR2101S). Phase current sensing for motoring quadrant operation is achieved by placing a sense resistor in the bottom Q2 source net and amplifying the signal using a low-cost op amp. However, in the laboratory prototype converter, a Hall effect current sensor was used for monitoring the actual phase current. A 100-μF 400-V electrolytic capacitor is suitably sized to ensure that the voltage ripple on the dc link is low and that long life is achievable given the operating capacitor current at peak operating power. The ac input stage includes a DS08 bridge rectifier, soft-start negative coefficient resistor, EMC filter, and system protection fuse. The control supply voltages are obtained from a printed circuit board mounted discrete linear power supply on the prototype, although a smaller and cheaper solution would be to use a switched-mode power supply circuit implemented with low-power high-voltage gate driver technology such as the LinkSwitch from Power Integration Inc. The control requirements for basic operation involve synchronizing phase energization with the rotor position. For this single-phase machine, a basic 8-bit microcontroller can be used to implement the control with very few additional external components. A single Hall sensor on the motor provides the necessary positional information, while the two pulse-width modulation peripheral pins provide the gate drive signals. An input capture peripheral is used to acquire the period of the Hall sensor signal, and the motor speed can be computed in a software for the speed control loop. The control can adjust the turn-on and turn-off gate firing angular positions. The peak current limit is set by an additional output pulse stream of variable duty which is filtered to provide the reference to an external current comparator. This provides full control in low-speed current chopping mode by turning off Fig. 15. Comparison of the measured and simulated phase currents at different speeds. (a) 1500 r/min. (b) 2000 r/min. a one gate firing signal when the current reaches the desired value causing the power converter to operate in freewheeling mode. A simulation model of the drive system (converter + motor) was constructed in Simulink, where the motor was modeled from the measured flux linkage versus current curves at different rotor positions using a lookup table. The simulated and measured phase currents at two different speeds with a fan load are shown in Fig. 15, where it can be observed that good agreements are achieved. V. STEADY-STATE MEASUREMENT RESULTS For such a small power drive system, the measurement of shaft power and torque could be strongly affected by some mechanical problems, e.g., shaft misalignment, the use of couplings between the motor and the torque transducer, and between the torque transducer and the load, which may cause extra vibrations and oscillations. A previous experiment using an optical rotary torque transducer (full-scale torque at 1 Nm) has shown that reliable steadystate torque measurement is difficult to achieve, due to the difficulty in aligning three separate shafts (motor, transducer, load) and two couplings. The measurement of this drive system was carried out by using the following procedure. 1) First, the input power to a dc motor at different speeds with free shaft was measured. Its resistance was measured, and the copper loss was calculated.

7 320 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 47, NO. 1, JANUARY/FEBRUARY 2011 VI. CONCLUSION In this paper, a new single-phase HSR motor has been proposed for low-cost pump applications. This motor has higher torque density than conventional single-phase SR motors by beneficially using PM flux, in a similar manner to the Cyrano motor [16]. However, to achieve high cogging torque in the zero-current period and to have high PM flux linkage, a special flux concentration design with cheap ferrite magnets is proposed. The rotor has a special design for reducing the hydraulic loss associated with wet running. The proposed motor is a suitable candidate to compete commercially with existing motors for low-cost pump applications. Fig. 16. Power and loss measurements for the HSR motor drive system using a fan as the load. 2) The dc motor was then connected to the HSR prototype motor. The input power to the dc motor at different speeds with the HSR motor open-circuited was measured. By comparing with the measured input power at the first step and subtracting the increment of dc motor copper loss, the mechanical loss of the HSR motor at different speeds may then be estimated (The additional core loss in the dc motor caused by the increased dc current was neglected; the core loss in the HSR motor caused by the magnets was neglected). 3) A fan was used as a load to the dc motor (no HSR motor connected), and the input power to the dc motor was measured at different speeds. By following the similar procedure described in the second step, the fan power as a function of speed may then be found. 4) The fan was connected to the HSR motor, and the HSR motor at different speeds was operated. The input power to the HSR was measured. Because the value of the resistance of the HSR motor is known and the phase current can be measured, the copper loss can be calculated. The fan power and mechanical loss characteristic over the speed range is known, so the input power minus the copper loss, the fan power and the mechanical loss, gives the core loss in relation to the armature current, which is the predominant core loss component. The power measurement results at different speeds are shown in Fig. 16. It could be observed that the mechanical loss, which is mainly the bearings loss (two standard bearings with an inner diameter of 10 mm and an outer diameter of 26 mm), is much higher than usual. This was later identified to be due to a misplacement of one of the bearings. At the highest speed that was recorded, the power-converter loss is around 5 W, and the copper plus core loss is around 13 W. The fan power and the mechanical loss are the motor output power, which in total is 28.6 W. This gives the drive system efficiency (motor + converter) of 62%. It should be noticed that this prototype motor has a degraded performance compared to the designed performance. As mentioned in Section III, it needs 30% more current to produce the desired torque, which greatly increases the copper loss and also the power-converter loss. REFERENCES [1] R. Krishnan and S. Lee, PM brushless DC motor drive with a new powerconverter topology, IEEE Trans. Ind. Appl., vol. 33, no. 4, pp , Jul./Aug [2] R. Krishnan, A novel single-switch-per-phase converter topology for four-quadrant PM brushless DC motor drive, IEEE Trans. Ind. Appl., vol. 33, no. 5, pp , Sep./Oct [3] R. Muller, Collector-less DC motor, U.S. Patent , Mar. 25, [4] C. Koechli, Y. Perriard, and M. Jufer, One phase brushless DC motor analysis, in Proc. Int. Conf. Elect. Mach., 1998, pp [5] A. Horng, Non-brush DC motor with an improved stator, U.S. Patent , Jan. 22, [6] A. Horng, Non-brush DC motor with new improved stator, U.S. Patent , Mar. 3, [7] J. Kim and R. Krishnan, Single-controllable-switch-based switched reluctance motor drive for low-cost variable-speed applications, in Proc. ECCE, San Jose, CA, Sep , 2009, pp [8] J. Kim and R. Krishnan, Novel two-switch-based switched reluctance motor drive for low-cost high-volume applications, IEEE Trans. Ind. Appl., vol. 45, no. 4, pp , Jul./Aug [9] K. Ha, C. Lee, J. Kim, R. Krishnan, and S.-G. Oh, Design and development of low-cost and high-efficiency variable-speed drive system with switched reluctance motor, IEEE Trans. Ind. Appl., vol. 43, no. 3, pp , May/Jun [10] J. C. Compter, Single-phase reluctance motor, U.S. Patent , Oct. 7, [11] G. E. Horst, Hybrid single-phase variable reluctance motor, U.S. Patent , Jun. 16, [12] G. E. Horst, Shifted pole single phase variable reluctance motor, U.S. Patent , Mar. 15, [13] P. Lurkens, Single-phase reluctance motor adapted to start in a desired direction of rotation, U.S. Patent , Jun. 27, [14] T. Higuchi, J. O. Fiedler, and R. W. De Doncker, On the design of a single-phase switched reluctance motor, in Proc. IEEE Int. Conf. Elect. Mach. Drives, Jun. 1 4, 2003, vol. 1, pp [15] J. M. Stephenson, Switched reluctance motors, U.S. Patent , Aug. 20, [16] V. Torok and K. Loreth, The world s simplest motor for variable speed control? The Cyrano motor, a PM-biased SR-motor of high torque density, in Proc. 5th Eur. Conf. Power Electron. Appl., Sep , 1993, vol. 6, pp Kaiyuan Lu received the B.S. and M.S. degrees from Zhejiang University, Hangzhou, China, in 1997 and 2000, respectively, and the Ph.D. degree from Aalborg University, Aalborg, Denmark, in He became an Assistant Professor in the Department of Energy Technology, Aalborg University, in 2005, where, since 2008, he has been an Associate Professor. His research interests include the design of permanent-magnet machines, finiteelement model analysis, and control of permanentmagnet machines.

8 LU et al.: NEW LOW-COST HSR MOTOR FOR ADJUSTABLE-SPEED PUMP APPLICATIONS 321 Peter Omand Rasmussen was born in Aarhus, Denmark, in He received the M.Sc.E.E. and Ph.D. degrees from Aalborg University, Aalborg, Denmark, in 1995 and 2001, respectively. In 1998, he became an Assistant Professor, and in 2002, he became an Associate Professor at Aalborg University. His research areas include magnetic gears and the design and control of switched reluctance and permanent-magnet machines. Steve J. Watkins (M 95) received the B.Sc., M.Phil., and Ph.D. degrees from the University of Leeds, Leeds, U.K., in 1982, the University of Huddersfield, Huddersfield, U.K., in 1988, and the University of Leeds in 2006, respectively. He was with a number of companies since first graduating in 1982, including Farnell Instruments Ltd. and Advanced Power Supplies Ltd., U.K., where he was engaged in the development of switchedmode power supplies. From 1988 to 1997, he was with Switched Reluctance Drives Ltd., U.K., where he was involved in the research and development of switched reluctance motor drive technology. In recent years, he has been with Fleadh Electronics Ltd., Leeds, U.K., as a Consultant specializing in the design and development of a variety of power electronics systems including grid-tie inverters, renewable energy generators, and electric vehicle motor drives. In 2005, he was a Visiting Professor at the Institute of Energy Technology, Aalborg University, Aalborg, Denmark. His technical interests include multilevel power converters, grid-tie inverters, switched reluctance/permanent-magnet brushless motor drives, and generators. He is the author of over ten journal and conference papers and is the holder of three patents. Frede Blaabjerg (S 90 M 91 SM 97 F 03) was born in Erslev, Denmark, on May 6, He received the M.Sc.EE. and Ph.D. degrees from Aalborg University, Aalborg, Denmark, in 1987 and 1995, respectively. He was with ABB-Scandia, Randers, Denmark, from 1987 to In 2000, he was a Visiting Professor at the University of Padua, Padua, Italy, as well as a part-time Program Research Leader in wind turbines with Research Center Risoe. In 2002, he was a Visiting Professor at Curtin University of Technology, Perth, Australia. He is currently a Full Professor of power electronics and drives and the Dean of the Faculty of Engineering and Science, Aalborg University. He is involved in more than 15 research projects with industry, including the Danfoss Professor Program in Power Electronics and Drives. His recent research interests include power electronics, static power converters, ac drives, switched reluctance drives, modeling, characterization of power semiconductor devices and simulation, power quality, wind turbines, and green power inverters. He is the author or coauthor of more than 350 publications in his research fields, including the book Control in Power Electronics (Academic, 2002). Dr. Blaabjerg was the recipient of the 1995 Angelos Award for his contributions to modulation technique and control of electric drives, the Annual Teacher Prize from Aalborg University in 1995, the Outstanding Young Power Electronics Engineer Award from the IEEE Power Electronics Society in 1998, five IEEE Prize Paper Awards during the last six years, the C. Y. O Connor Fellowship from Perth, Australia, in 2002, the Statoil Prize for his contributions to power electronics in 2003, and the Grundfos Prize in acknowledgment of his international scientific research in power electronics in He is an Associate Editor for the Journal of Power Electronics and for Elteknik. He has been an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, and he is currently the Editor-in-Chief of the IEEE TRANSACTIONS ON POWER ELECTRONICS. He is a member of the European Power Electronics and Drives Association, the IEEE Industry Applications Society Industrial Drives, Industrial Power Converter, and Power Electronics Devices and Components Committees, and the IEEE Industry Applications Society. He served as a member of the Danish Technical Research Council in Denmark from 1997 to 2003, and from 2001 to 2003, he was its Chairman. He has also been the Chairman of the Danish Small Satellite Program and the Center Contract Committee. He became a member of the Danish Academy of Technical Science in 2001, and in 2003, he became a member of its Academic Council. From 2002 to 2003, he was a member of the Board of the Danish Research Councils. In 2004, he became the Chairman of the Program Committee on Energy and Environment.