D 12 T 711 D 37 L 2. DC link capacitors. C 3. Capacitor in parallel with the battery. C dc. DC link equivalent capacitance. V dc

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1 2018 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 A Bidirectional High-Power-Quality Grid Interface With a Novel Bidirectional Noninverted Buck Boost Converter for PHEVs Omer C. Onar, Member, IEEE, Jonathan Kobayashi, Student Member, IEEE, Dylan C. Erb, Student Member, IEEE, and Alireza Khaligh, Senior Member, IEEE Abstract Plug-in hybrid electric vehicles (PHEVs) will play a vital role in future sustainable transportation systems due to their potential in terms of energy security, decreased environmental impact, improved fuel economy, and better performance. Moreover, new regulations have been established to improve the collective gas mileage, cut greenhouse gas emissions, and reduce dependence on foreign oil. This paper primarily focuses on two major thrust areas of PHEVs. First, it introduces a grid-friendly bidirectional alternating current/direct current ac/dc dc/ac rectifier/inverter for facilitating vehicle-to-grid (V2G) integration of PHEVs. Second, it presents an integrated bidirectional noninverted buck boost converter that interfaces the energy storage device of the PHEV to the dc link in both grid-connected and driving modes. The proposed bidirectional converter has minimal grid-level disruptions in terms of power factor and total harmonic distortion, with less switching noise. The integrated bidirectional dc/dc converter assists the grid interface converter to track the charge/discharge power of the PHEV battery. In addition, while driving, the dc/dc converter provides a regulated dc link voltage to the motor drive and captures the braking energy during regenerative braking. Index Terms Alternating current/direct current (ac/dc) dc/ac grid interface converter, bidirectional converters, noninverted buck boost dc/dc converter, plug-in hybrid electric vehicles (PHEVs), vehicle-to-grid (V2G). NOMENCLATURE V s Alternating current (AC) grid voltage. L 1 Alternating current/direct current (AC/DC) DC/AC converter inductor. T 16 AC/DC DC/AC converter insulated gate bipolar transistors (IGBTs). Manuscript received June 18, 2011; revised October 17, 2011, January 11, 2012, and March 3, 2012; accepted March 19, Date of publication March 31, 2012; date of current version June 12, This work was supported in part by the U.S. National Science Foundation under Grant , Grant , and Grant The review of this paper was coordinated by Prof. F. Assadian. O. C. Onar is with the Energy and Transportation Division, Oak Ridge National Laboratory, Oak Ridge, TN USA. J. Kobayashi is with the Department of Mechanical Engineering, University of California, Berkeley, CA USA. D. C. Erb is with the Department of Mechanical Engineering and also with the MIT Energy Initiative, Massachusetts Institute of Technology (MIT), Cambridge, MA USA. A. Khaligh is with the Power Electronics, Energy Harvesting, and Renewable Energies Laboratory, Department of Electrical and Computer Engineering, University of Maryland, College Park, MD USA. Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /TVT D 12 T 711 D 37 L 2 AC/DC DC/AC converter diodes. Bidirectional DC/DC converter IGBTs. Bidirectional DC/DC converter diodes. Bidirectional DC/DC converter inductor. V ab Voltage across the grid and L1. C 1 2 DC link capacitors. C 3 Capacitor in parallel with the battery. C dc DC link equivalent capacitance. V dc DC link voltage for both plug-in and driving modes. V batt Battery voltage. V peak Peak value of the grid voltage. V C1, V C2 C1 and C2 capacitor voltages. Iref Reference grid current (peak). I P Reference grid current (instantaneous). x, y, z Combinational logic Boolean variables. P Batt Measured charge/discharge power. Pref Reference charge/discharge power. D Duty cycle for DC/DC converter switching. R Triangular carrier waveform. G Gate signals. Vdc,ref Reference DC link voltage. T (s) DC/DC converter voltage loop controller transfer function. I batt Battery current. I batt,ref Reference battery current. f s Switching oscillator frequency. THD Total harmonic distortion. SoC State of charge (of the battery). v ce (θ), V CE(sat) On-saturation voltage for IGBT. v F (θ),v F Forward voltage drop for the diode. E on (θ) Turn-ON commutation loss. E off (θ) Turn-OFF commutation loss. E D_rev(θ) Diode reverse recovery loss. η Efficiency. η Change in efficiency. P in Input power. P o Output power. P loss Power loss. GHG Greenhouse gas emission. V2G Vehicle to grid. ISO Independent system operator. nth switch. T n /$ IEEE

2 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2019 I. INTRODUCTION PLUG-IN electric vehicle (PHEV) technology is the most promising candidate for meeting the future s ever-growing transportation needs. PHEVs introduce reduced fuel consumption, higher performance, and lower emission [1] [3]. By the time the next generations of PHEVs are brought to market, certain issues will need to be addressed. One imperative issue is the method by which these vehicles will be recharged and if the current grid can sustain the increased demand due to more PHEVs. The additional charging load of the PHEVs will bring in new load levels, patterns, and load characteristics, particularly the PHEV battery charger s load characteristics [4]. There are some studies with regard to the load level increasing impacts of PHEV penetration in power systems [5] [8]. However, the effects of PHEV chargers are not only limited to the increased load level but may also affect the power quality in terms of reactive power and harmonic distortions [9] [14] if conventional chargers are employed. The negative effects of the harmonics include transmission, distribution, and transformer overloading; additional losses on power system; decreased power system stability; increased skin effect losses; equipment, appliance, or device failures or damages; protection system malfunctioning or failure; insufficient reactive power compensation due to increase in frequency; resonant effects; interfering with communication and phone lines; and control and communication system malfunctioning. According to the U.S. Department of Energy Alternative Fuels and Advanced Vehicles Data Center, hybrid electric vehicles (HEVs) had been sold in the U.S. as of March The Department of Energy s target for electric vehicles (EVs) and PHEVs is 1 million vehicles by When PHEVs achieve a large market share, the grid could suffer if the PHEVs charging is always unidirectional, is uncoordinated with poor power factor, and draws distorted currents [15], [16]. Note that PHEVs have the capability of representing a large energy source for the grid. Tapping this resource could eliminate grid issues such as the constant need to provide load and generation balancing, frequency regulation, transmission congestion, time-of-use demand charges, and the need for voltage regulation, power quality, and renewables integration, which are simply a result of the demand variations that occur every day in addition to the constant need for voltage and frequency regulation [17]. To perform grid-connected vehicle battery applications, an advanced power electronic grid interface that can provide V2G bidirectional power flow with high power quality is required. An advanced interface system can respond to the charge/discharge commands that were received from an ISO, an aggregator, or a utility to provide demand-side management based on the needs of the power systems. This advanced interface should accomplish this task without causing any reactive power or power quality issues to the grid. If this V2G-enabled interface tracks the reference charge/discharge power, coordinated charging would be achieved, which could significantly reduce the potential gridrelated problems of PHEVs [18], [19]. In the near-future smart grid environment, ISOs, aggregator agencies, or utilities may determine the reference charge/discharge power by considering the drivers requirements, the SOC of the batteries, and the state of the grid. Smart grid communication infrastructure is also expected to command the residential appliances or commercial/industrial loads to provide regulation or spinning reserves through the demand response programs. A comprehensive review of the bidirectional converters have been presented in [20] with respect to their advantages, drawbacks, and other aspects. A comprehensive analysis of power converters employed in offboard chargers have been detailed in [21], including a basic voltage source inverter bridge, a VIENNA rectifier, a phase-shifted full-bridge converter, a half-bridge inductor-inductor-capacitor (LLC) converter, a three-phase interleaved converter, and the phase-shifted ZVS full-bridge converter. An onboard charger for PHEV applications that consists of a diode-bridge rectifier, followed by an interleaved boost converter with power factor correction (PFC), a high-frequency inverter and a transformer, and a diode-bridge rectifier that is cascaded to a battery filter circuit is proposed in [22]. The proposed topology achieves efficiency up to 94%, a power factor that ranges from 0.97 to 0.995, and a THD that ranges from 4% to 24%. References [23] and [24] present the energy efficiency in PHEV chargers along with the evaluation and comparison of frontend ac/dc topologies. Evaluated converter topologies include the conventional PFC boost converter (diode-bridge rectifier, followed by a boost converter), interleaved PFC boost converter, bridgeless PFC boost converter, phase-shifted semibridgeless PFC boost converter [25], bridgeless interleaved boost converter [26], and bridgeless interleaved resonant PFC boost converter [27]. Among these converters, the semibridgeless boost converter produces 5% 43% at 240 V input at different load levels, and the bridgeless interleaved boost converter produces about 4% 42% THD at 240 V input at different load levels. At 120 V input, the THD of these two converters varies from 5% to 24% and from 4% to 14%, respectively. The three-phase bidirectional battery charger that was presented in [28] is a three-phase bridge inverter with a bidirectional dc/dc converter and results in % of THD. As identified in [9], a report by the California Energy Commission reported battery charger input current THD variation up to 28% over the charging cycle. Reference [9] also states that a Ford Escort onboard battery charger measured input current THD of 59.6% at an output current of 15.7 A. It is clear that high penetration of PHEV chargers in a distribution network could cause a significant increase in the system THD. For the bidirectional noninverted buck boost part of the proposed system, the comparison may include a conventional buck boost converter. The converters can step up or down the input voltage but cannot provide bidirectional power flow, and they require an inverting transformer, because their output voltage is negative with respect to the input voltage [29]. Although some topologies can be noninverted [30] [41], [30] [32] and [34] [41] employ more than one switch in the pulsewidth modulation (PWM) mode, resulting in higher switching losses. Among these topologies, although they provide buck or boost operations, bidirectional power flow cannot be achieved in the topologies in [30], [33], and [37]. The conventional

3 2020 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 two-quadrant bidirectional converters can operate the buck mode in one direction and the boost mode in the other direction; however, they cannot operate vice versa [33],[36]. Bidirectional power flow with bucking and boosting capabilities can be achieved in two cascaded two-quadrant bidirectional converters; however, more than one high current inductor is required in these topologies [31], [35]. Although two switches and two inductors are used in [36], only unidirectional bucking or boosting can be achieved. In the case of a dual-active bridge dc/dc converter, all switches are operated in the PWM mode; therefore, switching losses are four times higher in the halfbridge case or eight times higher in the full-bridge case than the proposed converter. Dual-active bridge dc/dc converters also require a transformer at the middle stage, which increases the overall losses, size, and cost [38] [41]. In [38], two inductors are required in addition to the transformer, and in [39], the number of inductors is three. In [40], bidirectional power flow is possible with ten switches and two inductors. Finally, in [42], the proposed dc/dc converter requires two transformers, with one being multiwinded, which may complicate the structure and add up to the cost, and it does not have the bidirectional operating capability. This paper presents a bidirectional grid interfacing power electronic converter that enables the beneficial V2G interactions while ensuring that all power that is delivered from or injected to the grid has a high power factor and negligible current harmonics. This combination of the multilevel (threelevel) bidirectional ac/dc converter with the proposed bidirectional dc/dc converter can accomplish these requirements while tracking the reference charge/discharge power. The dc/dc part is also employed to provide a regulated dc link voltage to the motor drive in typical driving conditions. The multilevel ac/dc converter has low device stress with a relatively smaller input inductor compared to conventional H-bridge counterparts and inherently higher waveform quality (low THD) over the other diode bridge or PWM rectifier/inverter-based applications [43], [44]. The noninverted operation capability of the proposed converter totally eliminates the need for an inverting transformer, which reduces the overall size and cost. The dc/dc converter part of the proposed system uses only one switch in the PWM mode; therefore, the controls are as simple as the conventional buck or boost dc/dc converters despite all the competences; furthermore, switching losses are not more than that of a conventional buck or boost dc/dc converters. The proposed grid interface topology can also be scaled to three-phase applications, although it is originally a single-phase converter. To achieve a three-phase multilevel converter, one more leg (or bridge arm) should be added in parallel to the existing twophase legs. In a three-phase application, each phase leg should have two clamping diodes, whereas the number of capacitors at the dc link and their connections stay the same. The proposed system, as shown in Fig. 1, has four different generalized operational modes. Two of these modes occur in the grid-connected mode to supply power to/from the battery from/to the grid. The other two modes occur while driving by supplying power from/to the battery to/from the dc link while accelerating or regenerative braking. Due to these features, this complete topology is an ideal candidate for meeting the inclu- Fig. 1. Proposed integrated converter with ac/dc (grid interface) and dc/dc (battery interface) converters for EV and PHEV applications. Fig. 2. AC/DC converter flow path in mode 1. sive needs of the PHEV industry and HEV to V2G-equipped PHEV conversions. This paper is organized as follows. In Section II, the topological overview and the operation modes are presented, along with the control systems. Simulation results for the proposed converter are given in Section III. Section IV focuses on the experimental results to evaluate and validate the capabilities of the proposed converter. Finally, the conclusions are drawn in Section V. II. SYSTEM DESCRIPTION AND OPERATING MODES The multilevel ac/dc converter and the bidirectional integrated dc/dc converter constitute the proposed topology shown on the respective left and right sides of Fig. 1. The bidirectional multilevel ac/dc converter is made up of components L 1, D 1 D 2, T 1 T 6 with their internal diodes, as well as C 1 C 2. A. Operational Modes of the Grid Interface Converter When the vehicle is plugged in, the ac/dc converter switches through six different modes using T 1 T 6. In mode 1, V dc is applied across the grid and input inductor by turning T 1, T 3, and T 6 ON, as shown in Fig. 2. During this interval, the voltages across both C 1 and C 2 simultaneously increase or decrease, and the current that was delivered to/from the grid decreases, because a negative voltage is applied on L 1. In mode 2, which is given in Fig. 3, T 2, T 4, and T 5 are turned ON to apply V dc across the grid and inductor to simultaneously charge or discharge C 1 and C 2. In this operation state, the grid current increases due to the positive voltage across L 1. That is, in this state, both the grid voltage and the applied voltage are negative.

4 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2021 Fig. 3. AC/DC converter flow path in mode 2. Fig. 5. AC/DC converter flow path in mode 4. Fig. 4. AC/DC converter flow path in mode 3. Fig. 6. AC/DC converter flow path in mode 5. Throughout mode 3, half of the dc link voltage or the C 2 voltage (V dc /2 = V C2 ) is applied across the grid and the inductor by switching ON T 3, T 4, and T 6, as shown in Fig. 4. Based on the instantaneous voltage of the grid and V C2, the current that is delivered to/from grid either increases or decreases. In this mode, if the capacitor C 2 is charged (rectifier mode), positive grid current flows through the grid, L 1, T 4, D 2, C 2, and T 4. If the capacitor C 2 is discharged (inverter mode), negative grid current flows through C 2, D 1, T 3, L 1, the grid, and T 6. These two different current flow paths are shown in Fig. 4, where the solid lines show the capacitor discharge (inverter mode), and the dashed lines show the capacitor recharge (rectifier mode). In mode 4, T 2, T 3, and T 4 are turned ON to apply negative half of the dc link voltage or the negative C 1 voltage ( V dc /2 = V C1 ) across the grid and the inductor, as shown in Fig. 5. According to the relationship between V s and V C1, the current that is delivered to/from grid increases or decreases. In this mode, if the capacitor C 1 is charged (rectifier mode), positive grid current flows through the grid, L 1, T 4, D 2, C 1, and T 2. If the capacitor C 1 is discharged (inverter mode), negative grid current flows through C 1, D 1, T 3, L 1, the grid, and T 2. These two different current flow paths are shown in Fig. 5, where the solid lines show the capacitor discharge (inverter mode), and the dashed lines show the capacitor recharge (rectifier mode). During mode 5, as shown in Fig. 6, the negative half-line cycle of V s is shorted across the inductor by keeping T 1, T 2, and T 3 ON. Therefore, the inductor current magnitude increases. In mode 6, T 4, T 5, and T 6 are turned ON, as shown in Fig. 7, so that the positive half-line cycle of V s can be shorted across the inductor; therefore, the inductor current increases once again. Fig. 7. AC/DC converter flow path in mode 6. The converter operation is provided by applying three different voltage levels V ab to the right side of the inductor, depending on the voltage that the grid applies to the inductor V s. Applying three-level voltages provides waveform shaping for the grid current; by switching between two different voltage levels with different pulsewidths, the current that is drawn from or injected to the grid can be controlled. The grid voltage variation and the applied V ab voltage levels over a period are provided in Fig. 8. Table I presents the gate-switching patterns for each mode and the corresponding V ab voltage applied across the inductor and the grid. The control strategy for both the ac/dc rectifying and dc/ac inverting is provided in Fig. 9, which generates the Boolean x, y, and z, which will determine the mode of operation and the switching signals for T 1 T 6. The controller requires four feedback interface inputs: grid voltage V s, grid current I s, and

5 2022 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 Fig. 8. Switching between modes with respect to V s. TABLE I SWITCHING PATTERNS FOR DIFFERENT MODES OF OPERATION dc link capacitor voltages V C1 and V C2. The peak of the V s voltage is selected to be the reference dc link voltage. Because V dc = V C1 + V C2, the peak of V s is compared with V C1 + V C2, and the error is fed through the voltage controller, which outputs the rough current reference. The grid voltage is normalized by dividing V s by V peak so that a unity sine wave in phase with V s is obtained. This unity sine wave is multiplied with the output of the voltage controller. To compensate for the imbalance between the dc link capacitors, their voltage difference is added up to the rough current reference to obtain the actual Iref.The actual grid current is then compared with Iref, and error is fed to the current proportional integral derivative controller. Consequently, three Boolean variables x, y, and z are generated based on the instantaneous state of the circuit. Variable x is determined by the current direction, variable y is determined by the ac voltage region detection, and the sign of the ac voltage determines the variable z. Finally, these variables are interpreted according to the combination given in Table II that represents the converter s mode of operation. B. Operational Modes for the Battery Interface Converter The proposed dc/dc converter can provide bidirectional power flow by bucking/boosting the dc link voltage or bucking/boosting the battery voltage. Leading automakers have introduced PHEVs that utilize battery packs with rated voltages lower than the dc link (motor drive) voltage but higher than the rectified ac voltage. For example, the Toyota Prius plug-in comes with a V battery pack, whereas the motor drive voltage is 650 V [45], and the vehicle can be charged with a 120 V standard North American wall outlet. Similarly, the Chevy Volt battery pack is 300 V, the motor drive voltage is V, and the battery can be recharged from 120 V supplies [46]. An add-on battery pack with lower voltage can also always be used in a retrofitted PHEV conversion [47]. Therefore, it is evident that a multifunctional dc/dc battery interface is required for PHEVs. The different conditions for plugged-in or driving modes are summarized in Table III. When the vehicle is plugged in, the rectified ac voltage (here, the dc link voltage) should be stepped up for grid charging, and the battery voltage should be stepped down for grid discharging. While the vehicle is driven, the battery voltage should be stepped up during acceleration, and the dc link voltage should be stepped down during regenerative braking or down hilling. The states of the switches with respect to these operational modes are mapped in Table IV. Modes 1 and 2: Plug-In Charging and Discharging: When charging the battery pack, the rectified ac voltage (dc link voltage) is usually maintained at the peak grid voltage. While stepping up the dc link voltage, D 3, T 7, L 2, T 9, D 5, and T 11 are used to form a boost converter, as shown in Fig. 10. In this mode, T 9 is the only switch that is operated in PWM. When T 9 is turned ON, as shown by the red dashed line, L 2 is energized by the dc link voltage through D 3, T 7, L 2, and T 9. When T 9 is turned OFF, both the inductor and the dc link supplies power to the battery, as shown in the solid and dashed red lines of Fig. 10. The battery voltage is stepped down with the buck converter that is made up of T 10, D 7, L 2, D 6, D 4, and T 8. In this mode, T 10 is operated in the PWM mode. When T 10 is ON, the battery current is delivered to the dc link while energizing the inductor through T 10, D 7 (as shown by the solid blue line), L 2, T 8, and D 4. When T 10 is turned OFF, the output current is recovered by the freewheeling diode D 6, decreasing the average current transferred from the battery to the dc link (as shown by the dashed blue lines). The current flow paths of this mode are presented in the blue lines in Fig. 10. Modes 3 and 4: Acceleration and Braking Modes for Driving: During driving, the battery voltage should be stepped up in acceleration or cruising conditions. A boost converter is formed by T 10, D 7, L 2, T 9, T 8, and D 4, as shown in Fig. 11. In this case, PWM switching signals are applied to T 9. When T 9 is turned ON (as shown by the dashed red lines), battery current flows through T 10, D 7, and T 9, and the inductor is energized. When T 9 is turned OFF, D 4 is forward biased, and the inductor current flows to the output, as shown in the solid red lines in Fig. 11. During regenerative braking, the motor drive inverter recovers braking energy; therefore, the dc link voltage increases. When a rise is detected in the dc link voltage, the converter should be operated in the buck mode from the dc link to the battery. A buck converter can be formed by D 3, T 7, D 6, L 2, D 5, and T 11, as shown in Fig. 11. In this condition, T 7 is in the PWM mode. When T 7 is turned ON, current from the dc link passes through D 3, T 7 (as shown by solid blue lines), L 2, D 5, and T 11 while energizing the inductor. When T 7 is OFF, the output current is freewheeled through D 6, D 5, and T 11 (as shown by solid blue lines), decreasing the average current transferred to the battery. The current flow path for this operation is shown the blue lines of Fig. 11. As shown in Figs. 12 and 13, all operations of the battery interface converter are combinations of buck-and-boost operations. For the proposed system, two different controllers are incorporated: one controller is for modes 1 and 2 (plug-in charging/discharging), and the other controller is for modes 3

6 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2023 Fig. 9. Block diagram of the ac/dc dc/ac converter controller. TABLE II SEQUENCE OF OPERATIONS AND THE COMBINATIONAL LOGIC TABLE III SEQUENCE OF OPERATIONS AND THE COMBINATIONAL LOGIC TABLE IV SEQUENCE OF OPERATIONS AND THE COMBINATIONAL LOGIC Fig. 11. Boost and buck power flows for acceleration/braking. Fig. 12. DC/DC converter s charging/discharging power controller. Fig. 13. DC/DC converter s cascaded controller for the driving mode. Fig. 10. Boost and buck power flows for plug-in charging/discharging. and 4 (acceleration/deceleration during driving). In the driving mode, it is important to provide a regulated dc link voltage to the motor drive, whereas in the plug-in mode, it is desired to control the charging or discharging power of the battery. Therefore, a power controller is used for plug-in modes, and a double-loop voltage and a current controller is employed for acceleration/braking modes. The battery charge/discharge power controller, as shown in Fig. 12, allows for tracking the reference charge or discharge power of the battery. Based on the SOC of the battery, user requirements, and the state of the grid, this reference power can be determined. If a fast charge is desired, the reference power Pref can simply be increased to the desired value, because the proposed converter can dynamically adapt to any changes in reference charge/discharge power. In the case of an application where constant current charge is followed by a constant voltage charge, the power interpretation of such charging can be implemented by accordingly varying the power. In constant current charging, the battery voltage would gradually increase, resulting in a ramping-up power rate, whereas in constant voltage, charging the current would gradually decrease, resulting in

7 2024 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 a ramping-down power rate. By applying these power rates as the reference power, both the constant current and the constant voltage charging can be achieved. During driving, the high-voltage bus is maintained at the reference motor drive inverter voltage by the cascaded voltage and current controller, as shown in Fig. 13. This enables discharging of the battery to the dc link during acceleration and regenerative recharging of the battery from the dc link while maintaining the proper voltage level for the hybrid vehicle. C. Efficiency Analysis Approach for the Grid Interface and Battery Interface Converter For the grid interface converter part of the proposed system, semiconductor power losses can be estimated by implementing v sat (θ) I l (θ) and E(θ) I l (θ) presented in the manufacturer datasheet of each device. Here, v sat (θ) is the ON-state saturation voltage, which is v ce (θ) for the insulated gate bipolar transistor (IGBT) and v F (θ) for the diode, whereas E(θ) is the energy losses in one commutation, where E on (θ) is the turn- ON commutation loss, E off (θ) is the turn-off commutation, and E D_rev(θ) is the diode reverse recovery commutation loss. For the loss calculations, simple models of these losses as functions of the carried current can be created as v ce (I l (θ)), v F (I l (θ)), E on (I l (θ)), E off (I l (θ)), and E D_rev(I l (θ)) for the devices. The conduction and the switching power losses can be calculated based on the models for each semiconductor device of the grid interface converter. The sum of the conduction and switching losses would give the total losses of the converter. Because the switching losses should be obtained by implementing every turn-on and turn-off instants during one reference period, turn turn-on, turn-off, and reverse recovery losses can be given by P on = 1 T Eon (I l (θ)) (1) P off = 1 T Eoff (I l (θ)) (2) P D_rev = 1 T ED_rev (I l (θ)). (3) Hence, the total switching losses are equal to the sum of the turn-on, turn-off, and reverse recovery losses, which is expressed by P Switching = P on + P off + P D_rev. (4) Here, note that there is not a certain switching frequency for the grid-side converter (the switching frequency is totally random, i.e., the converter does the switching whenever necessary and transits between modes, depending on the grid voltage, grid current, dc link voltage, and the dc link capacitors voltage imbalance). Although there is not a certain switching frequency for the overall grid-side converter, we can obtain that the switching frequency of the power switches T 2 and T 6 is equal to the line frequency, which is 60 Hz (see the operational modes). On the other hand, when the converter switches between different levels (0 to V dc /2, V dc /2toV dc,0 to V dc /2, and V dc /2to V dc ), the switching frequency is also relatively low, i.e., around only a few kilohertz. Therefore, (1) (3) should be computed with different reference periods for different devices. For example, switches T 2 and T 6 would have much less switching losses compared with other switches for the same amount of time. The conduction losses in the proposed grid interface converter occur when a semiconductor device is ON and there is conducting current. The conduction losses for an IGBT is given by (5), as shown below, whereas (6) expresses the conduction losses for a diode, as shown below. Therefore, the total losses can be calculated by (7), as shown below. P cond,sw = 1 2π P cond,d = 1 2π 2π 0 2π 0 v sat (θ).i l (θ) dθ (5) v F (θ).i l (θ) dθ (6) P cond,t otal = P cond,sw + P cond,d. (7) Finally, the total losses will be equal to the sum of the switching and the conduction losses as P Total_loss = P Switching + P cond,t otal. (8) Because the switching losses of the grid-side converter are relatively lower, the conduction losses of the proposed converter are slightly higher than the H-bridge conventional inverters. However, H-bridge conventional converters switch at much faster switching frequency, resulting in their switching losses being higher than the proposed converter. Therefore, we can obtain that the overall losses of both the proposed grid interface and the conventional H-bridge inverters can be in close proximity. Equations (1) (8) are implemented in a MATLAB script program along with the current variation for a given period time while taking the switching periods of individual devices into account. Because the efficiency can be described as a function of the input power and the losses, the efficiency of the converter can be calculated by η = 1 P Total_loss P in (9) Therefore, the efficiency of the proposed grid interface converter is obtained as a function of the percentage of the input power, as given in Fig. 14. The peak efficiency of the proposed grid interface is 95.25%. The rated power is 18 kw based on the switch ratings of 600 V and 30 A. The switching losses of the proposed bidirectional noninverted buck boost converter are very similar to a regular noninverting buck boost converter. To make the criteria of comparison clear, the compared converter should have noninverted output and a relatively wider voltage for both the battery and the dc link. Even if the proposed converter is compared with the simplest fundamental buck or boost dc/dc converters for each of its modes, the switching losses are identical, because the proposed converter has only one switch in the PWM mode in all of the modes. Because the number of switches that operate in PWM is the same for the proposed and the

8 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2025 TABLE V COMPARATIVE CHANGE IN EFFICIENCY OF THE DC/DC CONVERTER FOR DIFFERENT MODES Fig. 14. Grid interface converter s efficiency. conventional converters, we can obtain that there is no increase in switching losses. However, it can be stated that the proposed converter has relatively slightly more conduction losses in all the operating modes. The additional conduction loss is mainly due to the additional switches or diodes in the current flow paths of the proposed converter. Because the proposed dc/dc converter has four different operating modes, losses should separately be examined, because all modes introduce different loss components. For example, in the plug-in charging mode, the dc/dc converter is operated in the boost mode from dc link to the battery. If the bottom line is a regular boost dc/dc converter, it can be found that the difference in loss is the conduction losses of a pair of an IGBT and a diode whereas the switching losses are the same. In dc analysis, the diode conduction loss would be P D = v F.I F, whereas the IGBT conduction loss would be P T = v CE(SAT).I CE. Therefore, in this mode, the change in losses can be expressed as ΔP loss,1 = P D3 + P T 7 + P T 11. (10) In the plug-in discharge mode, where the dc/dc converter is operated in the buck mode from the battery to the dc link, it is shown that the additional conduction losses compared to a conventional buck converter are due to a pair of additional diode (D 7 and D 4 ) and the switch (T 8 ). Therefore, in this mode, additional conduction losses can be calculated by ΔP loss,2 = P D7 + P D4 + P T 8. (11) For the acceleration in the driving mode, the dc/dc converter is operated in the boost mode from the battery to the dc link; therefore, the additional conduction losses compared to a conventional boost converter are due to a pair of additional conduction switches (T 10 and T 8 ) and the diode (D 7 ).This results in additional losses to be ΔP loss,3 = P T 10 + P T 8 + P D7. (12) Similarly, for the regenerative braking in the driving mode, the dc/dc converter is operated in the buck mode from the dc link to the battery; therefore, additional conduction losses occur due to the forward biased diodes D 3 and D 5 and the conducting switch T 11, as given by ΔP loss,4 = P T 11 + P D3 + P D5. (13) To estimate the comparative change in efficiency, η is identified as the efficiency of the conventional buck or boost mode, and η is defined as the efficiency of the proposed converter with additional losses. If P o is the output power and the P in is the input power, the change in efficiency can be obtained as Δη = η η = P o P o. (14) P in P in +ΔP loss The comparative change in efficiency for all of the four modes is formulated as a function of P in, P out, P loss, and ΔP loss ; ΔP loss for different modes is given by (10) (13). For these analyses, as used in the experiments, HGTG30N60A4D IGBT modules and FFPF30U60STTU power diodes from Fairchild Semiconductor are used, where the IGBT s V CE(SAT) is 1.6 V, whereas the diode s V F is 2.1 V, as given in the datasheets. Changes in efficiency values are summarized in Table V with respect to different operating modes. D. Cost Comparative Analysis for the Proposed Converter For the grid interface converter part of the proposed system, cost comparison analysis is based on the kilovolt-ampere ratings of the total power electronic semiconductors needed, and this total kilovolt-ampere ratings of the total semiconductor devices needed is compared with a conventional H-bridge inverter topology. In an H-bridge inverter topology, when one phase is in operation, there are two devices in series, and these two-series switches are subject to the entire V dc voltage. Assuming that V dc is 1 p.u., the voltage on per device would be 0.5 p.u. However, in the proposed multilevel grid interface converter, there are three semiconductor devices in series during the operation of one phase. Moreover, these three devices in series are subject to only half of the V dc voltage, because the converter switches between 0 and V dc /2, V dc /2 and V dc, 0 and V dc /2, and V dc /2 and V dc. Therefore, each device is subject to 1/3 the half dc link voltage. Two topologies are compared in Table VI in terms of their number of devices, voltage stress on the device, and the total kilovolt-ampere ratings in per unit. In these analyses, current is assumed to be constant and the same for two of the topologies, because it depends on the load, and the converters are analyzed under the same loading conditions. Table VI shows that the total kilovolt-ampere rating of the proposed converter is 1.32 p.u., whereas a conventional H-bridge inverter s total kilovolt-ampere rating is 2.0. p.u. Therefore, although a larger number of semiconductor switches are utilized in the proposed converter, their voltage rating can be less, and the proposed converter would be less expensive due to the reduced device ratings.

9 2026 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 TABLE VI COMPARATIVE KILOVOLT-AMPERE ANALYSES OF THE PROPOSED GRID INTERFACE CONVERTER AND A CONVENTIONAL H-BRIDGE INVERTER TABLE VIII SIMULATION CONDITIONS AND CIRCUIT PARAMETERS TABLE VII COMPARATIVE KILOVOLT-AMPERE ANALYSES OF THE PROPOSED DC/DC CONVERTERS WITH CONVENTIONAL APPROACHES However, it reduces the size, cost, and space requirements of the inductors by 50% while adding the grid discharge functionality. If a unidirectional charging scenario is required, the boost functionality of the dc/dc converter (from the dc link to the battery) would not be needed. In this case, the following two semiconductor devices can be eliminated: 1) D 3 and 2) T 7. However, there would not be any significant modification on the grid interface converter, because it is inherently bidirectional. However, the control strategy can be less complicated in the case of a unidirectional approach. For the dc/dc converter, to provide the same functionality, four dc/dc converters would be needed with conventional converters: two of these converters would be boost dc/dc converters (one converter for plug-in and one converter for the driving mode), and the other two converters would be buck dc/dc converters (one converter for plug-in and one converter for the driving mode). In this case, instead of one inductor, four inductors would be needed for each of the converters. However, commercially available EVs and PHEVs do not currently have the capability to inject power back to the grid. In addition, for the driving mode, they utilize a two-quadrant converter to provide both the boost and buck functions either for acceleration or regenerative braking modes. In Table VII, the proposed converter is compared with the conventional approach with four conventional dc/dc converters and the commercially available vehicle power electronics, although they do not have the plug-in discharge capability. As shown in Table VII, the proposed converter adds only two more semiconductor devices; however, it reduces the number of inductors from four to one compared with the two buck boost converter approach. Because the inductor core and winding materials are extremely more expensive than the semiconductor devices, it is always desirable to add two more semiconductor devices to reduce the number of inductors by three. Moreover, inductors would require much more space compared to the space requirement of two switches. Therefore, we can state that the proposed dc/dc converter would reduce both the cost and the size of the conventional approach for the same functionality basis. Compared with the commercial approach without the grid discharge mode, the proposed converter has six more semiconductor switches. III. SIMULATION ANALYSIS AND RESULTS Simulations for the grid interface converter provide the major waveforms and results for a typical application along with the comparisons in terms of noise emissions and harmonics related to the conventional grid interface converter and the proposed converter. Simulations for the dc/dc converter part provide how the dc/dc converter would work in a real drive cycle and how it successfully regulates the motor drive input voltage while managing the input and output power of the battery. Simulations are performed using MATLAB, Simulink, SimPowerSystems, Signal Processing Toolbox, and Control System Toolbox products. For the simulations, the charging conditions of a Toyota Prius PHEV are considered in the proposed system. The specifications of the simulation model are summarized in Table VIII. The variations of the ac voltage, ac current, and V ab voltage across the grid and inductor are given in Fig. 15 for 1800-W reference charging power. As shown in Fig. 15, the current drawn from the grid is in phase with the grid voltage, resulting in a unity power factor. Moreover, the grid current has near-zero harmonic distortion and has an almost-perfect sinusoidal shape. To test the dynamic performance of the reference charge power tracking feature of the proposed system, a step change in reference power has been applied at t=5 from 1000 W to 1800 W. The voltage that is applied to the battery, the dc link power that is transferred to the battery, the battery current, the battery SoC, and the grid current variations are provided in Fig. 16. In this mode, note that the rectified ac voltage (dc link voltage) is boosted more than the battery s rated voltage of V to charge the battery. To increase the charging power from 1000 W to

10 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2027 Fig. 15. Simulation results for the grid-charging mode. Fig. 17. Simulation results for the grid-discharging mode. Fig. 16. Simulation results for the grid-charging mode. (a) Battery charging voltage. (b) Battery charging current. (c) DC link power. (d) SoC of the battery. (e) Current drawn from the grid W, the converter applies higher voltage to the battery terminals, as shown in Fig. 16(a); therefore, the battery charging current increases, as shown in Fig. 16(b). Fig. 16(c) shows the dc link power that is used to charge the battery through the dc/dc converter. The SoC increment of the battery is displayed in Fig. 16(d) to show how its rate of change differs with two different charging power levels. Because the charging power increases, the ac/dc converter draws more current from the grid [see Fig. 16(e)] to maintain the dc link at the grid s peak voltage. In the grid-connected discharging mode, the ac voltage, ac current, and V ab voltage are recorded, as shown in Fig. 17 for a 1000-W reference discharge rate. Similar to the gridcharging mode, the current that is injected to the grid is almost totally free of distortions, but it is in reverse phase (180 ), with the grid voltage indicating that the power is supplied back to the grid. Because the zero crossings of the ac voltage and current are identical, no reactive power is injected to the Fig. 18. Simulation results for the grid-discharging mode. (a) Battery discharge voltage. (b) Battery discharge current. (c) DC link power. (d) SoC of the battery. (e) Current injected to the grid. grid. A step change in the reference discharge power has been applied at t=5 from 800 W to 1500 W to test the dynamic performance of the reference discharge power tracking feature of the proposed system. The battery voltage during discharge, the dc link power that is injected to the grid, the battery current, the battery SoC, and the current, which is injected to the grid, are shown in Fig. 18. In the plug-in discharge mode, the battery voltage is stepped down to slightly more than the peak of the grid s voltage, and this voltage is inverted to the ac. Once the controller senses that the reference discharge power is changed, more current is drawn from the battery, as shown in Fig. 18(b), to meet this demand. Therefore, the battery terminal voltage drops, as shown in Fig. 18(a), with respect to the drawn current. Fig. 18(c) shows the dc link power that is transferred from the battery through the dc/dc converter. Fig. 18(d) shows the SoC decrease of the battery. When the reference discharge

11 2028 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 TABLE IX EXPERIMENTAL CONDITIONS AND CIRCUIT PARAMETERS the battery to the dc link, and the battery voltage is stepped up. When the load demand is negative (mode 4), the motor drive captures the braking energy, and the dc/dc converter is operated in the buck mode to recover the braking energy. As shown in Fig. 19(b), T 9 is in the PWM mode, whereas the load demand is positive, and T 7 is operated in the PWM mode, whereas the load demand is negative. The regulated dc link voltage at 650 V is provided in Fig. 19(c) while the vehicle is being driven. Fig. 19(d) presents the SoC of the battery, which decreases when the load demand is positive and slightly increases during regenerative braking. Fig. 19. Simulation results for the driving buck and boost modes. (a) Load current. (b) Switches in the PWM mode. (c) DC link voltage. (c) Battery s SoC. power increases, the SoC drops faster. The multilevel converter injects more current to the grid, as shown in Fig. 18(e), to keep the dc link regulated. For the second phase of the simulations, the dynamic performance of the battery interface converter is tested over a portion of a well-known Urban Dynamometer Drive Schedule (UDDS) drive cycle for t=[1200, 1369]. A Toyota Prius PHEV with V nominal battery voltage and 650 V nominal dc link (motor drive inverter input) voltage is considered. The load demand in this portion includes acceleration, regenerative braking, and idling conditions that may occur in a typical driving scheme. While the vehicle is driven, a regulated dc link voltage should be provided to the motor drive while supplying the load demands and recapturing the braking energy. In Fig. 19, the load current profile that corresponds to the drive cycle [see Fig. 9(a)], map of switches operated in PWM [see Fig. 9(b)], regulated dc link voltage variation [see Fig. 9(c)], and the SoC of the battery [see Fig. 9(d)] are provided. When the load demand is positive (mode 3), the vehicle is accelerating, cruising, or idling; therefore, the power flow direction is from IV. EXPERIMENTAL SETUP AND RESULTS The details of the experimental setup of the proposed converter are provided in Table IX. For the experimental validation, a smaller scale setup has been built in the Energy Harvesting and Renewable Energies Laboratory with 30-V rms ac voltage, 42 V dc link voltage, and 24 V battery voltage both in the grid-connected and highvoltage bus loading modes. Because the proposed topologies are new and have not been built or tested, it would be more appropriate to build the small-scale prototypes rather than the full-scale high-power converters. Moreover, due to safety purposes and to protect the students and the laboratory equipment, a smaller scale prototype with a lower voltage rating is preferred to serve as a proof of principle. Because the dc link voltage is higher than the battery voltage, the battery voltage is stepped up in grid discharging and loading conditions, whereas the dc link voltage is stepped down in gridcharging and regenerative tests. A picture of the experimental setup of the proposed topology is depicted in Fig. 20. As a feedback and control systems interface, the TMS320F2812 digital signal processing (DSP) module from Texas Instruments has been employed. For programming the DSP, processing the feedback signals, and control realizations, the Target Support Package 4.1 for the TMS320C2000 and Embedded IDE Link for Code Composer Studio from MathWorks Inc. have been used. These tools allow for deploying generated code onto the real-time embedded microcontrollers and DSPs. The experimental results of the grid connection mode are presented in Fig. 21. In Fig. 21(a), CH-1 is the ac grid

12 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2029 Fig. 20. Experimental setup of the proposed converter. Fig. 22. Experimental results for the V batt to V dc boost mode. Fig. 23. Experimental results for the V dc to V batt buck mode. Fig. 21. Experimental results for the grid-connected mode. (a) Grid-charging ac quantities and the dc link voltage. (b) Grid-charging battery voltage and current. (c) Grid-discharging quantities and the dc link voltage. voltage reduced with a transformer for small-scale testing, CH-2 is the V ab voltage across the input and the transformer, CH-3 is the current drawn from the grid, and CH-4 is the dc link voltage in the grid-charging mode, all in 20 V and 20 A per division scales. The dc link voltage (CH-1, 20 V/div), battery voltage (CH-2, 20 V/div), and battery current (2 A/div) are presented in Fig. 21(b) for grid charging. Finally, the ac voltage, V ab voltage, current that is injected to the grid, and dc link voltage are provided in Fig. 21(c) in CH-1 Ch-4, respectively. Again, the grid current is 180 out of phase with the ac input voltage, indicating that the power is supplied back to the grid. Because the battery voltage and current are similar in the grid-discharging mode, they are not included; the major difference is the direction of the battery

13 2030 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 61, NO. 5, JUNE 2012 Fig. 24. Harmonic content and THD comparisons of a (a) multilevel converter, (b) diode-bridge rectifier, and (c) full-bridge PWM converter. Fig. 25. Noise emissions of the multilevel (CH-1) and full-bridge PWM (CH-2) converters. (a) [0, F s/2] frequency band. (b) [0, F s] frequency band. current. The experimental results verify that the ac power quality is in compliance with the IEEE-1547 and 1519 standards, which deal with the harmonic levels and other power system quality aspects of the equipment. To verify the capabilities of the proposed system in the driving mode, the battery to the dc link boost mode and the dc link to the battery buck mode are tested. The experimental results of the V batt to V dc boost mode of operation (acceleration or cruising) are presented in Fig. 22, where CH-1 is V dc,ch-2 is V batt, CH-3 is the input current, and CH-4 is the gate signal of the switch T 9 operated in PWM. It can be observed in Fig. 22 that the proposed converter can provide regulated dc link voltage at its reference value by boosting 24 V V batt voltage to 42 V V dc output voltage. The experimental results for the V dc to V batt buck mode of operation (regenerative braking) is presented in Fig. 23, where CH-1 is V batt,ch-2isv dc, CH-3 is input current, and CH-4 is the PWM switching signals of the switch T 7. In Fig. 23, it is shown that the input voltage of V dc is stepped down to about 24 V of the battery terminal voltage. The THD percentages of the conventional diode bridge rectifier, a conventional PWM converter, and the presented multilevel converter are shown in Fig. 24, including all the harmonic components. In addition, the switching noise that is emitted by the PWM converter and the multilevel converter is analyzed and compared in Fig. 25. This figure presents the results of the spectrum analyzer. According to Figs. 24 and 25, the multilevel topology exhibits higher performance in terms of power quality and noise emissions compared with the other grid interface converters. These results show that the proposed converter resulted in higher power quality by improving the waveforms and reducing the THDs from 61.02% to 2.35% compared with the diode bridge rectifier and from 4.71% to the 2.35% compared to the full-bridge PWM rectifier. At 100 khz, the proposed topology reduces the electromagnetic interference (EMI) noise from 41 db to 71 db compared to the full-bridge PWM-based power electronic interface at the [0,F s /2] frequency band due to the improved power quality. At the [0,F s ] frequency band, the proposed converter further reduces the EMI emissions from 83 db to 119 db due to the improved power quality. V. C ONCLUSION Because more PHEVs will be on the roads in the near future, it is important to consider the effects that large numbers of plugin vehicles might have on the grid. To avoid these issues, a highpower-quality grid interface must handle the energy exchange between the vehicle and the grid with minimal current harmonic distortions, high power factor, and less noise. The presented grid interface enables V2G interactions, which could improve the efficiency of the grid. The proposed dc/dc converter, similar to the grid interface converter, exhibits excellent performance both in the grid-connected and driving modes of operation. This converter can step down and step up the dc link and battery voltage, providing bidirectional power flow. Efficiency analysis and

14 ONAR et al.: GRID INTERFACE WITH BIDIRECTIONAL NONINVERTED BUCK BOOST CONVERTER FOR PHEVs 2031 the kilovolt-ampere-ratings-based cost comparative analysis are included to represent the feasibility of the proposed converter. The proposed grid interface converter reduces the THD of an H-bridge converter from 4.71% to 2.35% and reduces the diode bridge rectifier boost converter combination s THD from 61.02% to 2.35%. The noise emission level of the proposed multilevel converter is 77 db, whereas the noise emission level of the H-bridge inverter was 41 db, which shows that the proposed multilevel converter causes less EMI effects. The kilovolt-ampere analysis of the grid interface converter shows that the proposed grid interface converter reduces the total kilovolt-ampere rating of a conventional H-bridge inverter by 66%. 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Frank, Power conversion, Auto Electron., pp. 8 24, Nov./Dec [Online]. Available: autoelectronics.com/mag/0711aef1.pdf [49] S. Schulz, Power electronics for electric and hybrid vehicles, in Proc. IEEE APEC, Austin, TX, Feb. 2008, pp Jonathan Kobayashi (S 08) received the B.Sc. degree in electrical engineering from Illinois Institute of Technology (IIT), Chicago, in He is currently working toward the M.Sc. degree with the Department of Mechanical Engineering, University of California, Berkeley. He was a Power Electronics Researcher with the Energy Harvesting and Renewable Energies Laboratory, Electric Power and Power Electronics Center, IIT, where he worked on experimental bidirectional ac/dc and dc/dc power electronic converters for plugin hybrid electric vehicles. From August 2009 to December 2009, he was a Teaching Assistant for the robotics lab course. From August 2008 to May 2009, he was with the Formula Hybrid Team, IIT, where he worked on control systems and data acquisition. Earlier, he was with the Hawaiian Electric Company, Honolulu, HI, where he worked on various projects related to transmission and distribution substations, system protection, and communications. He is an expert in computer, web, and electrical power technologies. Mr. Kobayashi received the First Hawaiian Bank Scholarship in , the Heald Scholarship, and the First Robotics Scholarship in He was included on the IIT Dean s List in Fall 2007 and Fall Dylan C. Erb (S 07) received the B.S. degree in general engineering from the University of Illinois, Urbana-Champaign, in He is currently working toward the M.S. degree in mechanical engineering with the Department of Mechanical Engineering, Massachusetts Intitute of Technology (MIT), Cambridge. He is currently with the Field Intelligence Laboratory, MIT, under the supervision of Prof. S. Sarma. His research interests include the optimization of energy storage packs for hybrid and electric vehicles. Omer C. Onar (S 05 M 10) received the B.Sc. and M.Sc. degrees in electrical engineering from Yildiz Technical University, Istanbul, Turkey, in 2004 and 2006, respectively, and the Ph.D. degree in electrical engineering from Illinois Institute of Technology, Chicago, in July He is currently with the Energy and Transportation Science Division, Oak Ridge National Laboratory, U.S. Department of Energy, Oak Ridge, TN. He is the author or a coauthor of more than 40 publications including journals, conference proceedings, and books. His research interests include power electronics; energy harvesting/scavenging; renewable energies, particularly solar, wind, and ocean energy conversion systems; grid interconnection of renewable energy sources; power management for sustainable energy systems; and electric/hybrid electric/plugin hybrid electric vehicles. Dr. Onar is a member of the IEEE Power and Energy Society, the IEEE Vehicular Technology Society, the IEEE Industrial Electronics Society, and the IEEE Power Electronics Society. He is the recipient of the Transportation Electronics Fellowship from the IEEE Vehicular Technology Society, the Joseph J. Suozzi INTELEC Fellowship in Power Electronics from the IEEE Power Electronics Society in 2009, and the distinguished Alvin M. Weinberg Fellowship from the Oak Ridge National Laboratory in July Alireza Khaligh (S 04 M 06 SM 09) received the B.S. and M.S. degrees in electrical engineering from Sharif University of Technology, Tehran, Iran, and the Ph.D. degree in electrical engineering from Illinois Institute of Technology (IIT), Chicago. He was a Postdoctoral Research Associate with the Department of Electrical and Computer Engineering, University of Illinois, Urbana-Champaign. He was an Assistant Professor with IIT. He is currently an Assistant Professor and the Director of the Power Electronics, Energy Harvesting, and Renewable Energies Laboratory (PEHREL), Department of Electrical and Computer Engineering, University of Maryland, College Park (UMCP). He is the author or a coauthor of more than 100 journal articles and conference proceedings. His research interests include the modeling, analysis, design, and control of power electronic converters, electric and plug-in hybrid electric vehicles, biomechanical energy harvesting, renewable energies, grid integration of distributed energy systems, and smart grid. Dr. Khaligh is the recipient of the Ralph R. Teetor Educational Award from the Society of Automotive Engineers in 2010, the Armour College of Engineering Excellence in Teaching Award from IIT in 2009, and the Exceptional Talents Fellowship and the Distinguished Undergraduate Student Award from Sharif University of Technology. He was the Program Chair of the 2011 IEEE Vehicle Power and Propulsion Conference (VPPC) and the Grants and Awards Chair of the 2012 IEEE Applied Power Electronics Conference and Exposition (APEC). He is a Program Cochair of the 2012 IEEE Transportation Electrification Conference and Expo (ITEC). He is an Associate Editor for the IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY (TVT) and a Guest Editor for the IEEE TVT Special Section on Sustainable Transportation Systems and the IEEE TRANSACTIONS ON POWER ELECTRONICS Special Issue on Transportation Electrification and Vehicle Systems.

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